System And Method For Common Mode Translation

ABSTRACT

System and method for common mode translation in continuous-time sigma-delta analog-to-digital converters. An embodiment includes a loop filter having an RC network coupled to a differential signal input, a Gm-C/Quantizer/DAC circuit (GQD) coupled to the loop filter, a common-mode level adjust circuit coupled to signal inputs of the GQD, and a tuning circuit coupled to the GQD and the common-mode level adjust circuit. The GQD evaluates an input signal provided by the RC network, computes a difference between a filtered input signal and the feedback quantization signal to generate an error signal, measures the error signal, and compensates for the error signal with sigma-delta noise-shaping. The common-mode level adjust circuit alters a common-mode level of a differential input signal to be substantially equal to a desired common-mode level and the tuning circuit provides a compensation voltage to the common-mode level adjust circuit based on a difference between the common-mode levels.

TECHNICAL FIELD

The present invention relates generally to a system and method forsignal processing, and more particularly to a system and method forcommon mode translation in continuous-time sigma-delta analog-to-digitalconverters.

BACKGROUND

A continuous-time sigma-delta analog-to-digital converter(continuous-time sigma-delta ADC) differs from a discrete-timesigma-delta ADC in that the continuous-time sigma-delta ADC makes use ofa loop filter while the discrete-time sigma-delta ADC uses aswitched-capacitor filter, which may require the use of fast settlingcircuits and an input buffer to eliminate sample glitches. Theswitched-capacitor filter may limit the signal bandwidth. Additionally,due to the thermal noise of the capacitors used in theswitched-capacitor filters, large capacitors may be needed to obtaingood signal-to-noise ratios.

The loop filter may have a topology that is active-Gm-C, active-RC, acombination of active-Gm-C and active-RC, or a′combination of active andpassive networks. A diagram shown in FIG. 1 illustrates a view of atypical prior art continuous-time sigma-delta ADC 100. Thecontinuous-time sigma-delta ADC 100 includes an input RC network 105 andan active-passive Gm-C/Quantizer/DAC circuit (GQD) 110.

The RC network 105, which may provide passive filtering of the inputsignals to the continuous-time sigma-delta ADC, may include resistors(R), such as resistors 155 and 156, and capacitors (C), such ascapacitors 160 and 161, for the positive and negative signal inputs tothe continuous-time sigma-delta ADC 100. The GQD 110 may include a loopfilter 170. a quantizer 175, and a feedback loop 180 from a positive anda negative output from the quantizer 175 back to the positive and thenegative inputs to the loop filter 170. Summing points combine thesignal from the respective feedback loop 180 and the respective inputsignal and provides it to the loop filter 170. The GQD 110 may evaluatean input signal (provided by the RC network 105), measure an errorsignal present in the input signal, and provide compensation for theerror signal. During normal operation of the GQD 110, a virtual shortcircuit may be maintained between the positive and the negative inputsof the loop filter 170 due to the GQD's high gain and its negativefeedback loop. The feedback loop 180 may include a digital-to-analogconverter (DAC) 185 to provide an analog version of the feedback of thequantizer 175 output.

Due to the nature of the GQD 110, the input common mode level of theloop filter 170 may be identical to the common mode level of the inputsignal. However, if the input signal is to be provided by a separateintegrated circuit (for example, an RF chip coupled to thecontinuous-time sigma-delta ADC 100), the common mode signal levels atthe input to the continuous-time sigma-delta ADC 100 could be too highor too low for proper operation and reliability. Therefore, there may bea need to accommodate different common mode levels at the input to thecontinuous-time sigma-delta ADC 100 to enable reliable and optimaloperation between the continuous-time sigma-delta ADC 100 and a varietyof RF chips. The common mode level may be higher than a supply voltageof the loop filter 170 in the GQD 110.

If the continuous-time sigma-delta ADC 100 is fabricated using alow-voltage process, reliability issues may arise due to the high commonmode level. Even with acceptable common mode levels, during start-up,overload conditions, or power supply loss, when the GQD 110 loop may beincapable of maintaining the summing junction (at the input to the loopfilter 170, for example) at the common mode level, the differentialswing of the input signal appears at the summing junction and may causea degradation in the reliability of the continuous-time sigma-delta ADC100.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by embodiments of thepresent invention which provide a system and a method for common modetranslation in continuous-time sigma-delta analog-to-digital converters.

In accordance with an embodiment, a continuous-time sigma-deltaanalog-to-digital converter (CT SD ADC) is provided. The continuous-timesigma-delta analog-to-digital converter includes a loop filter having aninput resistor-capacitor (RC) network coupled to a differential signalinput, a Gm-C/Quantizer/DAC circuit (GQD) coupled to the loop filter, acommon mode level adjust circuit coupled to signal inputs of the GQD,and a tuning circuit coupled to the GQD and to the common mode leveladjust circuit. The GQD evaluates an input signal provided by the inputRC network, computes a difference between a filtered input signal andthe feedback quantization signal to generate an error signal, measuresthe error signal, and compensates for the error signal with sigma-deltanoise-shaping. The common mode level, adjust circuit alters a commonmode level of a differential input signal to be substantially equal to adesired common mode level, and the tuning circuit provides acompensation voltage to the common mode level adjust circuit based on adifference between the common mode level of the differential inputsignal and the desired common mode level.

In accordance with another embodiment, a circuit for adjusting a commonmode level of a second circuit is provided. The circuit includes a firstcurrent supply coupled between a first input of the second circuit and apower rail, and a second current supply coupled between a second inputof the second circuit and the power rail. The first input and the secondinput make up a differential input, and the first current supply and thesecond current supply provide a current path between a respective inputand the power rail based on a control signal provided to the respectivecurrent supply.

In accordance with another embodiment, a method for tuning a circuit isprovided. The method includes determining a difference between a commonmode level of an input signal to the circuit and a desired common modelevel, generating a compensation voltage based on the difference, andapplying the compensation voltage.

An advantage of an embodiment is that implementation of the embodimentis simple and may be readily added to existing continuous-timesigma-delta analog-to-digital converters without significantmodification.

A further advantage of an embodiment is that relatively littleintegrated circuit real estate is required, helping to keep the cost ofthe integrated circuit low.

Yet another advantage of an embodiment is that the embodiment enablesthe tuning of the adjustments to the common mode level. This may allowthe use of the embodiment in a wide variety of applications, furtherenhancing its appeal.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiments disclosed may be readily utilized as a basisfor modifying or designing other structures or processes for carryingout the same purposes of the present invention. It should also berealized by those skilled in the art that such equivalent constructionsdo not depart from the spirit and scope of the invention as set forth inthe appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the embodiments, and the advantagesthereof, reference is now made to the following descriptions taken inconjunction with the accompanying drawings, in which:

FIG. 1 is a diagram of a typical continuous-time sigma-delta ADC;

FIG. 2 is a diagram of a prior art technique for providing common modelevel protection in a continuous-time sigma-delta ADC;

FIGS. 3 a through 3 c are diagrams of common mode adjust circuits;

FIG. 4 is a diagram of a schematic of an exemplary continuous-timesigma-delta ADC with a common mode adjust circuit;

FIG. 5 is a diagram of a schematic of an exemplary continuous-timesigma-delta ADC with a common mode adjust circuit; and

FIGS. 6 a through 6 c are diagrams of sequences of events used inadjusting a common mode adjust circuit of an exemplary continuous-timesigma-delta ADC.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the embodiments are discussed in detail below.It should be appreciated, however, that the present invention providesmany applicable inventive concepts that can be embodied in a widevariety of specific contexts. The specific embodiments discussed aremerely illustrative of specific ways to make and use the invention, anddo not limit the scope of the invention.

The embodiments will be described in a specific context, namely acontinuous-time sigma-delta ADC. The invention may also be applied,however, to other integrated circuits wherein there is a desire toprovide common mode level protection, such as in a continuous timesigma-delta DAC, and so on.

With reference now to FIG. 2, there is shown a diagram illustrating acontinuous-time sigma-delta ADC 200, wherein the continuous-timesigma-delta ADC 200 includes a prior art technique for providing commonmode level protection. The common mode level protection comes in theform of a voltage clamp and a series switch 190 for each input to thecontinuous-time sigma-delta ADC 200. The voltage clamp and the seriesswitch 190 however, only provides high voltage protection withoutstepping down the common mode level to an optimum value (e.g., about thesame as the common mode level produced by the output of the GQD 110).Furthermore, the voltage clamp and the series switch 190 provides highvoltage protection at the expense of large area and potentiallysignificant signal distortion and clipping due to the presence of clampsand series switches in the signal path. Additionally, the voltage clampand the series switch 190 may not be able to be maintained in the eventof power supply loss.

With reference now to FIGS. 3 a through 3 c, there are shown diagramsillustrating embodiments of common mode level protection circuitry for acontinuous-time sigma-delta ADC. In a majority of situations, a commonmode level as provided to an input of the continuous-time sigma-deltaADC may be higher than a desired common mode level of thecontinuous-time sigma-delta ADC. Therefore, a current sink may be usedto pull the common mode level down. The diagram shown in FIG. 3 aillustrates a circuit 300 containing two current sinks (NMOS transistors310 and 311, for example). When the NMOS transistors 310 and 311 areturned on (as controlled by a control voltage (also referred to as abiasing voltage) “CONTROL VOLTAGE A” coupled to their gate terminals), acurrent path is created to ground and the common mode level of thepositive input signal is pulled down to a level determined by thevoltage drop across resistors coupled to the positive signal input(resistors 155 and 156 of the RC network 105), for example. Similarly,the common mode level of the negative signal input is pulled to a leveldetermined by a voltage drop across resistors coupled to the negativesignal input of the RC network 105.

The voltage drop may be determined by the value of the resistors 155 and156 and by the value of the drain current of NMOS transistors 310 and311. A first of the two NMOS transistors, for example NMOS transistor310, may be coupled to the positive signal input of a continuous-timesigma-delta ADC, and a second of the two NMOS transistors, for exampleNMOS transistor 311, may be coupled to the negative signal input of acontinuous-time sigma-delta ADC. Although shown to be NMOS transistors,other types of transistors, such as PMOS, BJT. DMOS, and so forth, maybe used with modification to the circuit 300. The illustration anddiscussion of NMOS transistors should not be construed as being limitingto either the scope or the spirit of the present invention.

Similarly, when the common mode level as provided to the input of thecontinuous-time sigma-delta ADC may be lower than the desired commonmode level of the continuous-time sigma-delta ADC, a current source maybe used to provide the current needed to pull the common mode level up.In this case the voltage rise at the positive signal input may bedetermined by the value of resistors 155 and 156 and the drain currentof either of PMOS pull-up transistors 330 and 331 that may be coupled tothe positive signal input as the diagram shown in FIG. 3 b illustrates.Similarly, the voltage rise at the negative signal input may bedetermined by the value of resistors coupled to the negative signalinput and either of the PMOS pull-up transistors 330 and 331 that may becoupled to the negative signal input. When the PMOS transistors 330 and331 are turned on (as controlled by a control voltage “CONTROL VOLTAGEB” coupled to their gate terminals), a current path is created to VDDand the common mode level is pulled up. Although shown to be PMOStransistors, other types of transistors, such as NMOS, BJT, DMOS, and soforth, may be used with modification to the circuit 320. Theillustration and discussion of PMOS transistors should not be construedas being limiting to either the scope or the spirit of the presentinvention.

The circuit(s) (circuit 300 and circuit 320 or a combination thereof)may be coupled to a continuous-time sigma-delta ADC at one of severallocations. A first location may be at the inputs to the continuous-timesigma-delta ADC, such as the continuous-time sigma-delta ADC 100, (shownas plane A in FIG. 1). If the circuit 300 (or circuit 320 or both) iscoupled to the continuous-time sigma-delta ADC 100 at the inputs of thecontinuous-time sigma-delta ADC 100, then an additional resistor mayhave to be added to each signal input of the continuous-time sigma-deltaADC 100. The resistor may be needed to produce a voltage drop necessaryto shift the common mode level. This may reduce the bandwidth of theinput filter (the RC network 105), consume valuable integrated circuitreal estate, and attenuate the input signal. Furthermore, thedifferential signal at the inputs of the continuous-time sigma-delta ADC100 is a large signal, which may make matching the circuit 300 verydifficult due to the finite output resistance of the current sinks.

A second location may be at the RC network 105 (shown as plane B in FIG.1). If the circuit 300 is coupled to the continuous-time sigma-delta ADC100 at the RC network 105, then the resistor 155 may be used to realizethe voltage drop needed to shift the common mode level. However, thevalue of the resistor 155 may typically be smaller than the value of theresistor 156 (normally the resistance of the resistor 155 is aboutone-half the resistance of the resistor 156), which may mean that thecircuit 300 may potentially need to contain high-current current sinks.The use of high-current current sinks to shift the common mode level mayconsume more power than necessary and may overload the common modefeedback circuit of an output stage of an RF circuit providing the inputsignal to the continuous-time sigma-delta ADC 100. Additionally, thedifferential signal at the RC network 105 may still be a large signal(the differential signal at the RC network 105 may be expressed as(resistor 156)/(resistor 155+resistor 156) of the differential signal atthe inputs of the continuous-time sigma-delta ADC 100). This may makematching the circuit 300 difficult and cascading may be necessary.

A third location may be the inputs to the GQD 110 (shown as plane C inFIG. 1). If the circuit 300 is coupled to the continuous-timesigma-delta ADC 100 at the GQD 110, then the resistors 155 and 156 maybe used to realize the voltage drop needed to shift the common modelevel of the positive signal input and corresponding resistors of the RCnetwork 105 may be used to realize the voltage drop needed to shift thecommon mode level of the negative signal input, implying that the valuesof the current sinks in the circuit 300 may be at a minimum.

Additionally, the differential signal at the inputs to the GQD 110 maybe small (due to the virtual short at the inputs to the GQD 110) due tothe loop operation. This may eliminate the need for any cascoding of thecircuit 300, yielding large headroom. In turn, this may allow for largeroverdrive (small transconductance) in the current sinks in the circuit300 and permit better matching and a negligible noise contribution.

The placement of the circuit 300 at the GQD 110 of the continuous-timesigma-delta ADC 100 may typically be perceived as a source of noiseperformance degradation for the continuous-time sigma-delta ADC 100since the circuit 300 is located at a summing junction, where thefeedback signal is added to the input signal. Since the noise from thecurrent sinks (NMOS transistors 310 and 311, for example) may not bedivided by any gain, a large noise may be added directly to the inputsignal without any scaling. However, since the current sinks may havelarge headroom, the current sinks may be designed with large overdrive(i.e., very small transconductance) to help minimize their noisecontribution. Furthermore, with wideband input signals, the noiseperformance of the continuous-time sigma-delta ADC 100 may be dominatedby quantization noise rather than flicker or thermal noise of theindividual circuit components. Therefore, the addition of the circuit300 to the continuous-time sigma-delta ADC 100 may have little impact onthe noise performance of the continuous-time sigma-delta ADC 100.

Since there is substantially no differential voltage swing present atthe input to the GQD 110, the current sinks added in the circuit 300 maynot introduce any distortion to the input signal. Any mismatch betweenthe current sinks may appear simply as a DC offset without anyharmonics. Additionally, since the circuit 300 does not require anyvoltage clamps or series switches, significant area (integrated circuitreal estate) may be saved and distortion problems associated withvoltage clamps and series switches are eliminated.

In an alternative embodiment, the diagram shown in FIG. 3 c illustratesa circuit 340 that includes two current sinks (NMOS transistors 310 and311, for example) and two protection circuits (PMOS transistors 350 and351, for example). As shown in FIG. 3 b, the two current sinks may bereplaced with current sources if there is a need to pull the common modelevel up to the desired common mode level instead of the need to pullthe common mode level down to the desired common mode level.Alternatively, the current sources may be added in addition to thecurrent sinks to provide both a pull up and a pull down capability toadjusting the common mode level. The two protection circuits, PMOStransistors 350 and 351, for example, may be used to protect the loopfilter 170 of the GQD 110 from any high voltage from the input signal inthe case of a power supply loss, wherein the current sinks, NMOStransistors 310 and 311, for example, will not be operational. The PMOStransistors 350 and 351 may be selected since PMOS devices with thecontrol voltage at their gates “CONTROL VOLTAGE C” may always be low inthe case of supply loss. With the control voltage low, the PMOS devicesare conducting, pulling the common mode level down towards circuitground. In normal operation, the protection circuits are turned off anddo not affect the operation of the GQD 110.

The two protection circuits, the PMOS transistors 350 and 351, forexample, may form a potential divider with the resistors 150 and 156 todrop the level of the input signal to a low level to protect the loopfilter 170. When the power supply is lost, the “CONTROL VOLTAGE C” maybe at ground potential and thus the protection provided by the twoprotection circuits may still be in effect. Although shown to be PMOStransistors, other types of transistors, such as NMOS, BJT, DMOS, and soforth, may be used with modification to the protection circuits. Theillustration and discussion of PMOS transistors should not be construedas being limiting to either the scope or the spirit of the presentinvention.

Although shown as single transistors, the current sinks (NMOStransistors 310 and 311, for example), the current sources (PMOStransistors 330 and 331, for example), and the protection circuits (PMOStransistors 350 and 351, for example) may be implemented using multipletransistors arranged in parallel if additional current sourcing andsinking capabilities are needed, with the number and size of thetransistors as needed to provide the required current handlingcapabilities, manufacturing process limitations, and so forth.

The current sinks in the circuit 300, NMOS transistors 310 and 311, forexample, may need to be tuned (adjusted). The control voltage “CONTROLVOLTAGE A” may need to be generated based on the input signal's commonmode level as well as the desired common mode level.

With reference now to FIG. 4, there is shown a diagram illustrating aschematic of a continuous-time sigma-delta ADC 400 with a tuning circuit405 for setting a control voltage used to tune current sinks to set acommon mode level. The continuous-time sigma-delta ADC 400 includes thecontinuous-time sigma-delta ADC 100 with the circuit 340 for common modelevel protection with additional supply loss protection. The tuningcircuit 405 may be coupled to the continuous-time sigma-delta ADC 400 atthe inputs to the GQD 110 like the circuit 340. The tuning circuit 405may compare both a positive input to the GQD 110 and a negative input tothe GQD 110 to a reference signal “VREF,” which may represent thedesired common mode level. The comparison between the positive input tothe GQD 110 and the reference signal may take place in a first pair oftransistors 410, while a second pair of transistors 415 may perform thecomparison between the negative input to the GQD 110 and the referencesignal. Current mirrors 420, 425, and 430 provide necessary current toassert a bias voltage to control the state of the current sinks(transistors 310 and 311, for example) in the circuit 340.

If there is no difference between the level of the positive input to theGQD 110 and the negative input to the GQD 110 (collectively, the commonmode level of the input signal) and the reference signal (the desiredcommon mode level), then the applied bias voltage goes to zero and thecurrent sinks of the circuit 340 are turned off. If there is a positivedifference between the level of the positive input to the GQD 110 andthe negative input to the GQD 110 (the common mode level of the inputsignal) and the reference signal (the desired common mode level), then apositive bias voltage is applied to the current sinks of the circuit 340and the current sinks are turned on and the inputs (both the positiveand the negative inputs) of the GQD 110 may be pulled down towards thedesired common mode level.

The first pair of transistors 410 and the second pair of transistors 415may become part of the capacitance needed at the input of the GQD 110and combine with capacitors in the RC network 105 to help reduce theoverall capacitance of the capacitors in the RC network 105, such as thecapacitor 161. The configuration as shown in FIG. 4 may have a reducedadditional integrated circuit real estate requirement that includes theremaining transistors in the current mirrors 420, 425, and 430.Additionally, since the capacitors in the RC network 105 are referencedto ground and the transistors in the first pair of transistors 410 andthe second pair of transistors 415 are reference to the supply (VDD),better capacitance linearity at the input of the GQD 110 may beachieved. The improved capacitance linearity may improve the overallperformance of the continuous-time sigma-delta ADC 400.

With reference now to FIG. 5, there is shown a diagram illustrating aschematic of a continuous-time sigma-delta ADC 500 with a tuning circuit505 for setting a control voltage used to tune current sinks to set acommon mode level. The tuning circuit 505 includes a diode connectedtransistor 510 that may be used to generate a bias voltage to controlthe state of the current sinks (transistors 310 and 311, for example) inthe circuit 340. The current sinks may be operated as current mirrors.

A reference current “IREF” of the diode connected transistor 510 may bedefined as VBG/RINT, where VBG is a band-gap voltage and RINT is aresistor similar (manufactured using the same manufacturing process) tothe resistors in the RC network 105, such as the resistors 155 and 156.With the resistor RINT being manufactured with the same manufacturingprocess as the resistors in the RC network 105, it may be ensured that avoltage drop across the resistors in the RC network 105, such as theresistors 155 and 156, may be accurately set by the band-gap voltage(VGB) and a ratio between the resistor RINT and a sum of the resistorsin the RC network 105 (resistors 155 and 156).

The setting of the voltage drop across the resistors in the RC network105 (a measure of programmability) may be implemented digitally throughswitching additional current sinks (similar to transistors 310 and 311)in the circuit 340 by a control bus “CONTROL.” Depending upon the valueof the voltage drop across the resistors in the RC network 105, a numberof current sinks may be turned on or turned off as needed. This mayrequire prior knowledge of the input signal's common mode level in orderto turn on the required number of current sinks.

However, since there is a limited number of unique RF integratedcircuits that may be attached to the continuous-time sigma-delta ADC 500and provide the input signals, it may be possible to determine a typicalcommon mode level for each unique RF integrated circuit and store themin a memory 515 of the continuous-time sigma-delta ADC 500. This mayoccur during manufacture of the continuous-time sigma-delta ADC 500 orit may occur during manufacture of a system containing thecontinuous-time sigma-delta ADC 500. The manufacturer may specify the RFintegrated circuit that may be coupled to the continuous-timesigma-delta ADC 500, and then based on a reference to the memory 515, acontrol circuit 520 may turn on a number of current sinks (via a controlbus 525) that may need to be turned on to properly set the voltage dropacross the resistors in the RC network 105.

With reference now to FIGS. 6 a through 6 c, there are shown diagramsillustrating sequences of events in adjusting the common mode level ofan input signal provided to a continuous-time sigma-delta ADC. Thediagram shown in FIG. 6 a illustrates a high-level sequence of events600 in adjusting the common mode level of an input signal provided to acontinuous-time sigma-delta ADC. The adjusting of the common mode levelmay begin with a determining of a difference between the common modelevel of the input signal and a desired common mode level (block 605).The difference between the common mode level and the desired common modelevel may then be used to generate a compensation voltage (block 610)that may be applied to a tuning circuit to bring the common mode levelto a level about equal to the desired common mode level (block 615).

The diagram shown in FIG. 6 b illustrates a sequence of events 630 usedin adjusting the common mode level of an input signal provided to acontinuous-time sigma-delta ADC, wherein a tuning circuit similar to thetuning circuit 405 is utilized to perform the adjusting. The sequence ofevents 630 may be an implementation of the sequence of events 600modified to meet the specific requirements of the tuning circuit 405.The tuning of the common mode level may begin with a comparison of theinput signals to the continuous-time sigma-delta ADC with the desiredcommon mode level (block 635). The comparison may be an implementationof the determining of the difference between the common mode level ofthe input signal and the desired common mode level (block 605). Thecomparison may be performed by the first pair of transistors 410 and thesecond pair of transistors 415, for example. Then, a bias voltage may begenerated based on the comparison of the common mode level and thedesired common mode level (block 640) and may be an implementation ofthe generating of the compensation voltage (block 610). Current mirrors420, 425, and 430 may be used to generate the compensation voltage, forexample. The bias voltage may then be provided to current sinks tochange the common mode level (block 645).

The diagram shown in FIG. 6 c illustrates a sequence of events 660 usedin adjusting the common mode level of an input signal provided to acontinuous-time sigma-delta ADC, wherein a tuning circuit similar to thetuning circuit 505 is utilized to perform the adjusting. The sequence ofevents 660 may be an implementation of the sequence of events 600modified to meet the specific requirements of the tuning circuit 505.The tuning of the common mode level may begin with the specifying of thetype and make of an integrated circuit, such as an RF chip, that may beproviding the input signal to the continuous-time sigma-delta ADC (block665). The specifying may be an implementation of the determining of thedifference between the common mode level of the input signal and thedesired common mode level (block 605) since each type and make ofintegrated circuit may be characterized by a typical common mode levelfor output signals provided. With the specified type and make ofintegrated circuit, it may be possible to retrieve a compensationvoltage, from a memory, for example (block 670) and may be animplementation of the generating of the compensation voltage (block610). With the compensation voltage, a number of current sinks may beturned on to provide a requisite voltage drop necessary to place acommon mode level at the input of a GQD that is substantially equal tothe desired common mode level (block 675). The voltage drop may berealized across the resistors that are part of an RC network present ina loop filter of the continuous-time sigma-delta ADC.

Although the embodiments and their advantages have been described indetail, it should be understood that various changes, substitutions andalterations can be made herein without departing from the spirit andscope of the invention as defined by the appended claims. Moreover, thescope of the present application is not intended to be limited to theparticular embodiments of the process, machine, manufacture, compositionof matter, means, methods and steps described in the specification. Asone of ordinary skill in the art will readily appreciate from thedisclosure of the present invention, processes, machines, manufacture,compositions of matter, means, methods, or steps, presently existing orlater to be developed, that perform substantially the same function orachieve substantially the same result as the corresponding embodimentsdescribed herein may be utilized according to the present invention.Accordingly, the appended claims are intended to include within theirscope such processes, machines, manufacture, compositions of matter,means, methods, or steps.

1-20. (canceled)
 21. A method of controlling a common mode input voltageat a differential input port of a feedback circuit, the feedback circuitcoupled to a differential system input through resistors, the methodcomprising: measuring the common mode input voltage at the differentialinput port; comparing the common mode input voltage at the differentialinput port to a reference voltage; and reducing a difference between thecommon mode input voltage at the differential input port and thereference voltage, the reducing comprising supplying a common mode errorcurrent to the differential input port.
 22. The method of claim 21,wherein the supplying comprises supplying the common mode error currentdirectly to the differential input port.
 23. The method of claim 21,wherein the supplying the common mode error current further comprises:generating a control voltage based on the difference between the commonmode input voltage and the reference voltage; and controlling aplurality of current sources coupled to the differential input port,wherein at least one of the plurality of current sources generates thecommon mode error current.
 24. The method of claim 21, wherein: thefeedback circuit comprises a Gm-C/Quantizer/DAC circuit for asigma-delta ADC; and the method further comprises operating theGm-C/Quantizer/DAC circuit.
 25. The method of claim 21, furthercomprising generating the reference voltage, the generating comprisinggenerating a band-gap voltage.
 26. The method of claim 21, furthercomprising clamping a voltage at the differential input port to reduceinput voltage excursions.